1. Field of the Invention
The present invention relates to an RFID transceiver device in an RFID system, and more particularly to an RFID transceiver device in which receiver noise is improved.
2. Description of the Related Art
As shown in FIG. 1, in an RFID system, a carrier signal is transmitted (P1) from an interrogator constituted by an RFID transceiver device 1 to a transponder for example an IC tag 2. The IC tag 2 modulates the received carrier signal with information data and sends this back to the RFID transceiver device 1 by reflection (back scattering). The RFID transceiver device 1 acquires information data by demodulating the reflected signal.
FIG. 2 shows an example of the configuration of an RFID transceiver device. This RFID transceiver device is connected to a data processing device, not shown, through an external interface I/F. A control and processing circuit 10 controls a local oscillation circuit 11 to generate a local oscillation signal corresponding to each channel.
The local oscillation signal that is generated from the local oscillation circuit 11 is modulated and amplified in power by a transmission circuit 12 before being emitted from an antenna 16 through a duplexer 13. The local oscillation signal is additionally supplied to a demodulation circuit in a reception circuit 14 and the demodulation circuit outputs information data by demodulating the reflected signal from the IC tag 2.
It is undesirable from the point of view of cost and size that separate antennas should be provided respectively for transmission and reception in FIG. 2, so a transceiving antenna 16 as shown in FIG. 2 may be employed for the RFID transceiver device 1.
Furthermore, since, in the case where the IC tag 2 is a passive tag, the operating power (power source energy) is obtained from the electromagnetic wave transmitted by the RFID transceiver device 1, the RFID transceiver device 1 needs to have large transmission power. In contrast, since the response transmission from the IC tag 2 is performed by reflection (back scattering), its power is very weak in comparison with the power of the electromagnetic wave transmitted by the RFID transceiver device. Thus, the RFID transceiver device 1 whose communications partner is a passive IC tag needs to have high output power in order to supply power source energy to the IC tag 2 and, at the same time, must be provided with a high sensitivity reception capability, since the back-scattered signal from the passive IC tag is very weak.
When a transceiving antenna 16 is employed, in order to isolate the transmission and reception signal, a duplexer (typically constituted by a circulator or coupler) 13 is provided; however, as mentioned above, the energy of the transmission signal is large, so leakage 15 of the transmission signal is generated, of a level that depends on the degree of isolation achieved by the duplexer 13. Also, as shown in FIG. 3, apart from the component 15a that passes through the duplexer with attenuation, the leakage components of the transmission signal include a component 15b that is reflected by the power feed terminal of the antenna 16. Also, as shown in FIG. 4, if the transmission and reception signal frequencies f1 and f2 are different, as they are in the case of for example a mobile telephone terminal, isolation of transmission and reception can be achieved by providing bandpass filters 12a, 14a as well as the duplexer 13. However, in the case of an RFID system, as shown in FIG. 3, the frequency of transmission (carrier signal) and reception (tag reflection signal) are the same, so isolation using filters is not possible. For the above reasons, phase noise of the carrier signal appears to be detected at the output of the demodulation circuit constituting the reception circuit 14. Also, due to the problem of saturation caused by the leakage of the transmission signal, low noise amplification cannot be performed upstream of the demodulation circuit.
The mechanism by which such phase noise is detected will further be described with reference to the drawings. FIG. 5 is an example of a specific configuration of the transmission circuit 12 and reception circuit 14 of the RFID transceiver device 1 shown in FIG. 2.
FIG. 6 is a view showing the input signal of the demodulation circuit 14b constituting the reception circuit 14. The input signals of the demodulation circuit 14b are the local oscillation signal 17 (FIG. 6A) from the local oscillation circuit 11 and the leakage component 15 (FIG. 6B) of the transmission signal including the component 15a that is transmitted, with attenuation, through the duplexer 13 and the reflected signal 15b from the antenna power feed terminal.
Consequently, assuming that the operation of the demodulation circuit 14b is multiplicative, when the higher order component is discarded, the output of the demodulation circuit 14b may be expressed by the expression (1).
                                          cos            ⁡                          [                                                ω                  ⁢                                                                          ⁢                  t                                +                                  p                  ⁡                                      [                    t                    ]                                                              ]                                ×                      cos            ⁡                          [                                                ω                  ⁡                                      (                                          t                      -                      τ                                        )                                                  +                                  P                  ⁡                                      [                                          t                      -                      τ                                        ]                                                              ]                                      ⇒                              1            2                    ⁢                      cos            ⁡                          [                                                ω                  ⁢                                                                          ⁢                  t                                +                                  P                  ⁡                                      [                    t                    ]                                                  -                                  P                  ⁡                                      [                                          t                      -                      τ                                        ]                                                              ]                                                          (        1        )            
The term that determines the magnitude of the phase noise component in the output of the demodulation circuit in expression (1) i.e.P[t]−P[t−τ]
is 0 when τ=0. In contrast, it increases with increasing τ if the phase noise is time-correlated.
However, in the above expression,cos[ωt+P[t]]
is the local oscillation signal (FIG. 6A) from the local oscillation circuit 11, andcos[ω(t−τ)+P[t−τ]]
is the leakage 15 (FIG. 6B) of the transmission signal.
On the other hand, the phase noise component is expressed by
      P    ⁡          [      t      ]        =            ∫      0      t        ⁢                  ⁢                  ⅆ        u            ⁢                        ∫          0          u                ⁢                              h            ⁡                          [                              u                -                v                            ]                                ⁢                      g            ⁡                          [              y              ]                                ⁢                                          ⁢                      ⅆ            v                              
whereg[t]
is the input noise of the VCO of the local oscillation circuit 11 andh[t]
is the frequency characteristics (loop filter characteristics) of the VCO input stage. Thus the phase noise component has a correlation with time.
From the above relationship, it can be seen that, if the paths to the output of the demodulation circuit 14b are respectively different for the local oscillation signal from the local oscillation circuit 11 and for the leakage 15 of the transmission signal, so that there is a time difference between the paths to the demodulation circuit as shown in FIG. 6, the correlation of the leakage 15 (FIG. 6B) of the transmission signal and the local oscillation signal (FIG. 6A) becomes smaller as the path time difference τ becomes larger: as a result, the noise component that is output from the demodulation circuit 14b also becomes larger. FIG. 7 is a graph showing the relationship between the path time difference and the noise level (relative value). From the graph of FIG. 7, it can be understood that the detected phase noise level becomes larger as the path time difference τ becomes larger and if there is no path time difference, the phase noise component is substantially cancelled out.
Laid-open Japanese Patent Application No. 2003-174388 may be mentioned as prior art. This Laid-open Japanese Patent Application No.2003-174388 mentions that the phase noise possessed by the carrier itself that is transmitted from the interrogator and the phase noise of the PLL oscillation circuit that is involved in synchronous detection appear in the demodulation signal and adversely affect reception sensitivity. The object of the invention set out in Laid-open Japanese Patent Application No. 2003-174388 referred to above is to prevent lowering of the reception sensitivity in synchronous detection by the interrogator.
However, the invention set out in Laid-open Japanese Patent Application No. 2003-174388 referred to above is an arrangement in which the phase of the local signal LO is corrected using the response signal from the tag as a reference. Such a configuration is effective in systems in which there is substantially no change in the amplitude/phase of the response signal and the leakage of the transmission signal i.e. systems in which the frequency is low, at about 13.56 MHz, and in which the distance to the transponder is small, at about 30 cm.
However, in the case where an RFID transceiver device and IC tag are employed with a distance of a few m in the UHF band (860 MHz to 960 MHz) or higher frequency bands, phase variations of 10 or more times 360° are experienced, depending on the distance.